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MIC4723
3A 2MHz Integrated Switch Buck Regulator
General Description
The Micrel MIC4723 is a high efficiency PWM buck (stepdown) regulator that provides up to 3A of output current. The MIC4723 operates at 2.0MHz and has proprietary internal compensation that allows a closed loop bandwidth of over 200KHz. The low on-resistance internal p-channel MOSFET of the MIC4723 allows efficiencies over 92%, reduces external components count and eliminates the need for an expensive current sense resistor. The MIC4723 operates from 2.7V to 5.5V input and the output can be adjusted down to 1V. The devices can operate with a maximum duty cycle of 100% for use in lowdropout conditions. The MIC4723 is available in the exposed pad 12-pin 3mm x 3mm MLF(R) package with a junction operating range from -40C to +125C. Data sheets and support documentation can be found on Micrel's web site at www.micrel.com.
Features
* * * * * * * * * * * * * 2.7 to 5.5V supply voltage 2.0MHz PWM mode Output current to 3A Up to 94% efficiency 100% maximum duty cycle Adjustable output voltage option down to 1V Ultra-fast transient response Ultra-small external components Stable with a 1H inductor and a 4.7F output capacitor Fully integrated 3A MOSFET switch Micropower shutdown Thermal shutdown and current limit protection Pb-free 12-pin 3mm x 3mm MLF(R) package -40C to +125C junction temperature range
Applications
* FPGA/DSP/ASIC applications * General point of load * Broadband communications * DVD/TV recorders * Point of sale * Printers/Scanners * Set top boxes * Computing peripherals * Video cards ___________________________________________________________________________________________________________
Typical Application
MIC4723 3.3V OUT Efficiency 96 4.5VIN 5VIN 94 92 90 5.5VIN 88 86 84 82 80 78 76 0 0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3A 2MHz Buck Regulator
3
MLF and MicroLeadFrame are registered trademarks of Amkor Technology, Inc. Micrel Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel +1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com
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Ordering Information
Part Number MIC4723YML Voltage Adj. Temperature Range -40 to +125C Package 12-Pin 3x3 MFL(R) Lead Finish Pb-Free
Pin Configuration
SW 1 VIN 2 PGND 3 SGND 4 BIAS 5 FB 6 EP 12 SW 11 VIN 10 PGND 9 PGOOD 8 EN 7 NC
12-Pin 3mm x 3mm MLF (ML)
Pin Description
Pin Number
1, 12 2, 11 3, 10 4 5 6 7 8 9 EP
Pin Name
SW VIN PGND SGND BIAS FB NC EN PGOOD GND
Pin Function
Switch (Output): Internal power P-Channel MOSFET output switch. Supply Voltage (Input): Supply voltage for the source of the internal P-channel MOSFET and driver. Requires bypass capacitor to GND. Power Ground. Provides the ground return path for the high-side drive current. Signal (Analog) Ground. Provides return path for control circuitry and internal reference. Internal circuit bias supply. Must be bypassed with a 0.1F ceramic capacitor to SGND. Feedback. Input to the error amplifier, connect to the external resistor divider network to set the output voltage. No Connect. Not internally connected to die. This pin can be tied to any other pin if desired. Enable (Input). Logic level low, will shutdown the device, reducing the current draw to less than 5A. Power Good. Open drain output that is pulled to ground when the output voltage is within 7.5% of the set regulation voltage. Connect to ground.
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Absolute Maximum Ratings(1)
Supply Voltage (VIN) .......................................................+6V Output Switch Voltage (VSW) ..........................................+6V Output Switch Current (ISW)............................................11A Logic Input Voltage (VEN) .................................. -0.3V to VIN Storage Temperature (Ts) .........................-60C to +150C
Operating Ratings(2)
Supply Voltage (VIN)..................................... +2.7V to +5.5V Logic Input Voltage (VEN) ....................................... 0V to VIN Junction Temperature (TJ) ........................ -40C to +125C Junction Thermal Resistance 3mm x 3mm MLF-12 (JA) .................................60C/W 3mm x 3mm MLF-12 (Jc) .................................28C/W
Electrical Characteristics(4)
VIN = VEN = 3.6V; L = 1H; COUT = 4.7F; TA = 25C, unless noted. Bold values indicate -40C< TJ < +125C.
Parameter
Supply Voltage Range Under-Voltage Lockout Threshold UVLO Hysteresis Quiescent Current Shutdown Current [Adjustable] Feedback Voltage FB pin input current Current Limit in PWM Mode Output Voltage Line Regulation Output Voltage Load Regulation Maximum Duty Cycle PWM Switch ONResistance Oscillator Frequency Enable Threshold Enable Hysteresis Enable Input Current Power Good Range Power Good Resistance Over-Temperature Shutdown Over-Temperature Hysteresis
Notes: 1. Exceeding the absolute maximum rating may damage the device. 2. The device is not guaranteed to function outside its operating rating. 3. Specification for packaged product only.
Condition
(turn-on)
Min 2.7 2.45
Typ
2.55 100
Max 5.5 2.65
Units
V V mV
VFB = 0.9 * VNOM (not switching) VEN = 0V
570 2
900 10 1.02
A A V nA A % % % %
2% (over temperature) ILOAD = 100mA
0.98
1
VFB = 0.9 * VNOM VOUT > 2V; VIN = VOUT+500mV to 5.5V; ILOAD= 100mA VOUT < 2V; VIN = 2.7V to 5.5V; ILOAD= 100mA 20mA < ILOAD < 3A VFB 0.4V ISW = 50mA; VFB = GND (High Side Switch)
3.5
5 0.07 0.2
100
95 200 300
m m MHz V mV A %
1.8 0.5
2 0.85 50 0.1 7
2.2 1.3 2.3
10
IPGOOD = 500A
150 160 25
250
C C
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Typical Characteristics
96 4.5VIN 94 92 90 88 86 84 82 80 78 76 0 0.5 1 1.5 2 2.5 OUTPUT CURRENT (A) 3 5.5VIN
MIC4723 3.3V OUT Efficiency
5VIN
94 92 90 88 86 84 82 80 0
MIC4723 2.5V OUT Efficiency
4.5VIN 5VIN
100 98 96 94 92 90 88 86 84 82 80 0
MIC4723 2.5VOUT Efficiency
3VIN 3.3VIN
5.5VIN
3.6VIN
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
90 88 4.5VIN 5VIN 86 84 82 80 5.5VIN 78 76 74 72 70 68 66 64 62 60 0.5 1 1.5 2 2.5 0 OUTPUT CURRENT (A)
MIC4723 1.8V OUT Efficiency
3
96 94 92 3VIN 3.3VIN 90 88 86 84 3.6VIN 82 80 78 76 74 72 70 68 66 0 0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
MIC4723 1.8V OUT Efficiency
3
86 84 82 80 78 76 74 72 70 68 66 64 0
MIC4723 1.5VOUT Efficiency
4.5VIN 5VIN
5.5VIN
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
92 90 3VIN 88 3.3VIN 86 84 82 3.6VIN 80 78 76 74 72 70 68 66 64 0 0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
MIC4723 1.5V OUT Efficiency
3
90 88 86 84 82 80 78 76 74 72 70 68 66 64 62 60 0
MIC4723 1.2V OUT Efficiency
4.5VIN 5VIN
5.5VIN
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
90 88 86 3VIN 84 3.3VIN 82 80 78 76 3.6VIN 74 72 70 68 66 64 62 60 0 0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
MIC4723 1.2VOUT Efficiency
3
78 76 74 72 70 68 66 64 62 60 0
MIC4723 1V OUT Efficiency
4.5VIN 5VIN
84 82 80 78 76 74 72 70 68 66
MIC4723 1V OUT Efficiency
OUTPUT VOLTAGE (V)
Load Regulation
1.010
3VIN
3.3VIN
1.005
5.5VIN
3.6VIN
1.000
0.995 3.3V VIN = 3.3V
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
64 0
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
0.990 0
0.5 1 1.5 2 2.5 OUTPUT CURRENT (A)
3
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Typical Characteristics (continue)
1.0010 1.0008 1.0006 1.0004 1.0002 1.0000 0.9998 0.9996 0.9994 0.9992 0.9990 2.7 3.2 3.7 4.2 4.7 5.2 SUPPLY VOLTAGE (V)
Line Regulation
1.010 1.008 1.006 1.004 1.002 1.000 0.998 0.996
Feedback Voltage vs. Temperature
2.5 2.4 2.3 2.2 2.1 2.0 1.9 1.8 1.7
Frequency vs. Temperature
0.994 0.992 V = 3.3V IN 0.990 20 40 60 80 TEMPERATURE (C)
1.6 V = 3.3V IN 1.5 20 40 60 80 TEMPERATURE (C)
1.2 1.0 0.8 0.6 0.4 0.2
Feedback Voltage vs. Supply Voltage
800 700 600 500 400 300 200
Quiescent Current vs. Supply Voltage
120 115 110 105 100 95 90 85 80
R DSON vs. Supply Voltage
VEN = VIN 0 0 1 2 3 4 5 SUPPLY VOLTAGE (V)
100 0 0
VEN = VIN 1 2 3 4 5 SUPPLY VOLTAGE (V) 6
75 70 2.7
3.2 3.7 4.2 4.7 5.2 SUPPLY VOLTAGE (V)
160 140 120 100 80 60 40 20 0
RDSON vs. Temperature
1.2 1.0 0.8 0.6 0.4 0.2
Enable Threshold vs. Supply Voltage
1.2 1.0 0.8 0.6 0.4 0.2
Enable Threshold vs. Temperature
VIN = 3.3V 20 40 60 80 TEMPERATURE (C) 0 2.7 3.2 3.7 4.2 4.7 SUPPLY VOLTAGE (V)
0
VIN = 3.3V 20 40 60 80 TEMPERATURE (C)
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Functional Characteristics
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Functional Diagram
VIN VIN P-Channel Current Limit BIAS
HSD
PWM Control
SW SW
EN
Enable and Control Logic
Bias, UVLO, Thermal Shutdown
Soft Start
FB
EA
1.0V
PGOOD 1.0V
SGND
PGND
MIC4723 Block Diagram
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Pin Description
VIN Two pins for VIN provide power to the source of the internal P-channel MOSFET along with the current limiting sensing. The VIN operating voltage range is from 2.7V to 5.5V. Due to the high switching speeds, a 10F capacitor is recommended close to VIN and the power ground (PGND) for each pin for bypassing. Please refer to layout recommendations. BIAS The bias (BIAS) provides power to the internal reference and control sections of the MIC4723. A 10 resistor from VIN to BIAS and a 0.1F from BIAS to SGND is required for clean operation. EN The enable pin provides a logic level control of the output. In the off state, supply current of the device is greatly reduced (typically <1A). Do not drive the enable pin above the supply voltage. FB The feedback pin (FB) provides the control path to control the output. For adjustable versions, a resistor divider connecting the feedback to the output is used to adjust the desired output voltage. The output voltage is calculated as follows:
R1 VOUT = VREF x + 1 R2 where VREF is equal to 1.0V. A feedforward capacitor is recommended for most designs using the adjustable output voltage option. To reduce current draw, a 10K feedback resistor is recommended from the output to the FB pin (R1). Also, a feedforward capacitor should be connected between the output and feedback (across R1). The large resistor value and the parasitic capacitance of the FB pin can cause a high frequency pole that can reduce the overall system phase margin. By placing a feedforward capacitor, these effects can be significantly reduced. Feedforward capacitance (CFF) can be calculated as follows:
SW The switch (SW) pin connects directly to the inductor and provides the switching current necessary to operate in PWM mode. Due to the high speed switching on this pin, the switch node should be routed away from sensitive nodes. This pin also connects to the cathode of the free-wheeling diode. PGOOD Power good is an open drain pull down that indicates when the output voltage has reached regulation. When power good is low, then the output voltage is within 10% of the set regulation voltage. For output voltages greater or less than 10%, the PGOOD pin is high. This should be connected to the input supply through a pull up resistor. A delay can be added by placing a capacitor from PGOOD to ground. PGND Power ground (PGND) is the ground path for the MOSFET drive current. The current loop for the power ground should be as small as possible and separate from the Signal ground (SGND) loop. Refer to the layout considerations for more details. SGND Signal ground (SGND) is the ground path for the biasing and control circuitry. The current loop for the signal ground should be separate from the power ground (PGND) loop. Refer to the layout considerations for more details.
C FF =
1 2 x R1 x 200kHz
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Application Information
The MIC4723 is a 3A PWM non-synchronous buck regulator. By switching an input voltage supply, and filtering the switched voltage through an Inductor and capacitor, a regulated DC voltage is obtained. Figure 1 shows a simplified example of a non-synchronous buck converter.
During the on-time, Figure 3 illustrates the high side switch is turned on, current flows from the input supply through the inductor and to the output. The inductor current is:
Figure 1. Example of non-synchronous buck converter
For a non-synchronous buck converter, there are two modes of operation; continuous and discontinuous. Continuous or discontinuous refer to the inductor current. If current is continuously flowing through the inductor throughout the switching cycle, it is in continuous operation. If the inductor current drops to zero during the off time, it is in discontinuous operation. Critically continuous is the point where any decrease in output current will cause it to enter discontinuous operation. The critically continuous load current can be calculated as follows;
2 V VOUT - OUT VIN IOUT = 2.0MHz x 2 x L
Figure 3. On-Time
charged at the rate;
(VIN - VOUT )
L To determine the total on-time, or time at which the inductor charges, the duty cycle needs to be calculated. The duty cycle can be calculated as; D= VOUT VIN
and the On time is;
Continuous or discontinuous operation determines how we calculate peak inductor current.
TON =
D 2.0MHz
Continuous Operation Figure 2 illustrates the switch voltage and inductor current during continuous operation.
Therefore, peak to peak ripple current is; Ipk -pk =
(VIN- VOUT ) x VOUT
VIN 2.0MHz x L
Since the average peak to peak current is equal to the load current. The actual peak (or highest current the inductor will see in a steady-state condition) is equal to the output current plus 1/2 the peak-to-peak current. Ipk = IOUT +
Figure 2. Continuous Operation
(VIN - VOUT ) x VOUT
VIN 2 x 2.0MHz x L
The output voltage is regulated by pulse width modulating (PWM) the switch voltage to the average required output voltage. The switching can be broken up into two cycles; On and Off.
Figure 4 demonstrates the off-time. During the off-time, the high-side internal P-channel MOSFET turns off. Since the current in the inductor has to discharge, the current flows through the free-wheeling Schottky diode to the output. In this case, the inductor discharge rate is
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Micrel, Inc. (where VD is the diode forward voltage); L The total off time can be calculated as; -
MIC4723
(VOUT
+ VD )
TOFF =
1- D 2.0MHz
When the inductor current (IL) has completely discharged, the voltage on the switch node rings at the frequency determined by the parasitic capacitance and the inductor value. In Figure 5, it is drawn as a DC voltage, but to see actual operation (with ringing) refer to the functional characteristics. Discontinuous mode of operation has the advantage over full PWM in that at light loads, the MIC4723 will skip pulses as nessasary, reducing gate drive losses, drastically improving light load efficiency.
Efficiency Considerations Calculating the efficiency is as simple as measuring power out and dividing it by the power in;
Efficiency = POUT x 100 PIN
Where input power (PIN) is; PIN = VIN x IIN and output power (POUT) is calculated as; POUT = VOUT x IOUT The Efficiency of the MIC4723 is determined by several factors.
Figure 4. Off-Time
* * *
Rdson (Internal P-channel Resistance) Diode conduction losses Inductor Conduction losses
Discontinuous Operation Discontinuous operation is when the inductor current discharges to zero during the off cycle. Figure 5 demonstrates the switch voltage and inductor currents during discontinuous operation.
* Switching losses Rdson losses are caused by the current flowing through the high side P-channel MOSFET. The amount of power loss can be approximated by;
PSW = R DSON x IOUT 2 x D
Where D is the duty cycle. Since the MIC4723 uses an internal P-channel MOSFET, Rdson losses are inversely proportional to supply voltage. Higher supply voltage yields a higher gate to source voltage, reducing the Rdson, reducing the MOSFET conduction losses. A graph showing typical Rdson vs input supply voltage can be found in the typical characteristics section of this datasheet. Diode conduction losses occur due to the forward voltage drop (VF) and the output current. Diode power losses can be approximated as follows; PD = VF x IOUT x (1 - D) For this reason, the Schottky diode is the rectifier of choice. Using the lowest forward voltage drop will help reduce diode conduction losses, and improve efficiency. Duty cycle, or the ratio of output voltage to input voltage, determines whether the dominant factor in conduction losses will be the internal MOSFET or the Schottky diode. Higher duty cycles place the power losses on the
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Figure 5. Discontinuous Operation
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Micrel, Inc. high side switch, and lower duty cycles place the power losses on the Schottky diode. Inductor conduction losses (PL) can be calculated by multiplying the DC resistance (DCR) times the square of the output current;
MIC4723
PL = DCR x IOUT 2 Also, be aware that there are additional core losses associated with switching current in an inductor. Since most inductor manufacturers do not give data on the type of material used, approximating core losses becomes very difficult, so verify inductor temperature rise. Switching losses occur twice each cycle, when the switch turns on and when the switch turns off. This is caused by a non-ideal world where switching transitions are not instantaneous, and neither are currents. Figure 6 demonstrates how switching losses due to the transitions dissipate power in the switch.
Figure 6. Switching Transition Losses
Normally, when the switch is on, the voltage across the switch is low (virtually zero) and the current through the switch is high. This equates to low power dissipation. When the switch is off, voltage across the switch is high and the current is zero, again with power dissipation being low. During the transitions, the voltage across the switch (VS-D) and the current through the switch (IS-D) are at middle, causing the transition to be the highest instantaneous power point. During continuous mode, these losses are the highest. Also, with higher load currents, these losses are higher. For discontinuous operation, the transition losses only occur during the "off" transition since the "on" transitions there is no current flow through the inductor.
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Component Selection
Input Capacitor A 10F ceramic is recommended on each VIN pin for bypassing. X5R or X7R dielectrics are recommended for the input capacitor. Y5V dielectrics lose most of their capacitance over temperature and are therefore, not recommended. Also, tantalum and electrolytic capacitors alone are not recommended due their reduced RMS current handling, reliability, and ESR increases. An additional 0.1F is recommended close to the VIN and PGND pins for high frequency filtering. Smaller case size capacitors are recommended due to their lower ESR and ESL. Please refer to layout recommendations for proper layout of the input capacitor. Output Capacitor The MIC4723 is designed for a 4.7F output capacitor. X5R or X7R dielectrics are recommended for the output capacitor. Y5V dielectrics lose most of their capacitance over temperature and are therefore not recommended. In addition to a 4.7F, a small 0.1F is recommended close to the load for high frequency filtering. Smaller case size capacitors are recommended due to there lower equivalent series ESR and ESL. The MIC4723 utilizes type III voltage mode internal compensation and utilizes an internal zero to compensate for the double pole roll off of the LC filter. For this reason, larger output capacitors can create instabilities. In cases where a 4.7F output capacitor is not sufficient, the MIC4723 offers the ability to externally control the compensation, allowing for a wide range of output capacitor types and values. Inductor Selection The MIC4723 is designed for use with a 1H inductor. Proper selection should ensure the inductor can handle the maximum average and peak currents required by the load. Maximum current ratings of the inductor are generally given in two methods; permissible DC current and saturation current. Permissible DC current can be rated either for a 40C temperature rise or a 10% to 20% loss in inductance. Ensure the inductor selected can handle the maximum operating current. When saturation current is specified, make sure that there is enough margin that the peak current will not saturate the inductor. Diode Selection Since the MIC4723 is non-synchronous, a free-wheeling diode is required for proper operation. A Schottky diode is recommended due to the low forward voltage drop and their fast reverse recovery time. The diode should be rated to be able to handle the average output current. Also, the reverse voltage rating of the diode should exceed the maximum input voltage. The lower the forward voltage drop of the diode the better the efficiency. Please refer to the layout recommendations to minimize switching noise. Feedback Resistors The feedback resistor set the output voltage by dividing down the output and sending it to the feedback pin. The feedback voltage is 1.0V. Calculating the set output voltage is as follows;
R1 + 1 VOUT = VFB R2
Where R1 is the resistor from VOUT to FB and R2 is the resistor from FB to GND. The recommended feedback resistor values for common output voltages are available in the bill of materials on page 19. Although the range of resistance for the FB resistors is very wide, R1 is recommended to be 10K. This minimizes the effect the parasitic capacitance of the FB node.
Feedforward Capacitor (CFF) A capacitor across the resistor from the output to the feedback pin (R1) is recommended for most designs. This capacitor can give a boost to phase margin and increase the bandwidth for transient response. Also, large values of feedforward capacitance can slow down the turn-on characteristics, reducing inrush current. For maximum phase boost, CFF can be calculated as follows;
C FF =
1 2 x 200kHz x R1
Bias filter A small 10 resistor is recommended from the input supply to the bias pin along with a small 0.1F ceramic capacitor from bias to ground. This will bypass the high frequency noise generated by the violent switching of high currents from reaching the internal reference and control circuitry. Tantalum and electrolytic capacitors are not recommended for the bias, these types of capacitors lose their ability to filter at high frequencies.
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Network Analyzer "R" Input Feedback Network Analyzer "A" Input Output
Loop Stability and Bode Analysis
Bode analysis is an excellent way to measure small signal stability and loop response in power supply designs. Bode analysis monitors gain and phase of a control loop. This is done by breaking the feedback loop and injecting a signal into the feedback node and comparing the injected signal to the output signal of the control loop. This will require a network analyzer to sweep the frequency and compare the injected signal to the output signal. The most common method of injection is the use of transformer. Figure 7 demonstrates how a transformer is used to inject a signal into the feedback network.
+8V
MIC922BC5
R1 1k
R3 1k
R4 1k 50
Network Analyzer Source
Figure 8. Op Amp Injection
Figure 7. Transformer Injection
A 50 resistor allows impedance matching from the network analyzer source. This method allows the DC loop to maintain regulation and allow the network analyzer to insert an AC signal on top of the DC voltage. The network analyzer will then sweep the source while monitoring A and R for an A/R measurement. While this is the most common method for measuring the gain and phase of a power supply, it does have significant limitations. First, to measure low frequency gain and phase, the transformer needs to be high in inductance. This makes frequencies <100Hz require an extremely large and expensive transformer. Conversely, it must be able to inject high frequencies. Transformers with these wide frequency ranges generally need to be custom made and are extremely expensive (usually in the tune of several hundred dollars!). By using an op-amp, cost and frequency limitations used by an injection transformer are completely eliminated. Figure 8 demonstrates using an op-amp in a summing amplifier configuration for signal injection.
R1 and R2 reduce the DC voltage from the output to the non-inverting input by half. The network analyzer is generally a 50 source. R1 and R2 also divide the AC signal sourced by the network analyzer by half. These two signals are "summed" together at half of their original input. The output is then gained up by 2 by R3 and R4 (the 50 is to balance the network analyzer's source impedance) and sent to the feedback signal. This essentially breaks the loop and injects the AC signal on top of the DC output voltage and sends it to the feedback. By monitoring the feedback "R" and output "A", gain and phase are measured. This method has no minimum frequency. Ensure that the bandwidth of the op-amp being used is much greater than the expected bandwidth of the power supplies control loop. An op-amp with >100MHz bandwidth is more than sufficient for most power supplies (which includes both linear and switching) and are more common and significantly cheaper than the injection transformers previously mentioned. The one disadvantage to using the op-amp injection method; is the supply voltages need to below the maximum operating voltage of the op-amp. Also, the maximum output voltage for driving 50 inputs using the MIC922 is 3V. For measuring higher output voltages, 1M input impedance is required for the A and R channels. Remember to always measure the output voltage with an oscilloscope to ensure the measurement is working properly. You should see a single sweeping sinusoidal waveform without distortion on the output. If there is distortion of the sinusoid, reduce the amplitude of the source signal. You could be overdriving the feedback causing a large signal response.
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Micrel, Inc. The following Bode analysis show the small signal loop stability of the MIC4723, it utilizes type III compensation. This is a dominant low frequency pole, followed by 2 zeros and finally the double pole of the inductor capacitor filter, creating a final 20dB/decade roll off. Bode analysis gives us a few important data points; speed of response (Gain Bandwidth or GBW) and loop stability. Loop speed or GBW determines the response time to a load transient. Faster response times yield smaller voltage deviations to load steps. Instability in a control loop occurs when there is gain and positive feedback. Phase margin is the measure of how stable the given system is. It is measured by determining how far the phase is from crossing zero when the gain is equal to 1 (0dB).
Bode Plot V =3.3V, V =1.8V, I
60 50 40 GAIN (dB) 30 20 10 L=1H 0 COUT = 4.7F GAIN
IN OUT
MIC4723 Being that the MIC4723 is non-synchronous; the regulator only has the ability to source current. This means that the regulator has to rely on the load to be able to sink current. This causes a non-linear response at light loads. The following plot shows the effects of the pole created by the nonlinearity of the output drive during light load (discontinuous) conditions.
Bode Plot VIN=3.3V,V OUT=1.8V,IOUT=50mA
60 50 40 GAIN (dB) 30 20
210 175 140
PHASE
70 35 0 -35 -70 -105 1M
10 L=1H 0 COUT = 4.7F -10 R1 = 10k GAIN R2 = 12.4k -20 C = 82pF FF -30 100 1k 10k 100k FREQUENCY (Hz)
OUT
=3A
210 175 140 70 35 0 -35 -70 -105 1M PHASE () 105
PHASE
3.3Vin, 1.8Vout Iout=50mA; * *
Phase Margin=90.5 Degrees GBW= 64.4KHz
-10 R1 = 10k R2 = 12.4k -20 C = 82pF FF -30 100 1k 10k 100k FREQUENCY (Hz)
Typically for 3.3Vin and 1.8Vout at 3A; * *
Phase Margin=47 Degrees GBW=156KHz
Feed Forward Capacitor The feedback resistors are a gain reduction block in the overall system response of the regulator. By placing a capacitor from the output to the feedback pin, high frequency signal can bypass the resistor divider, causing a gain increase up to unity gain.
Gain and Phase vs. Frequency
0 L=1H -1 C = 4.7F OUT -2 R1 = 10k -3 R2 = 12.4k -4 CFF = 82pF -5 -6 -7 -8 -9 -10 100 25 GAIN PHASE BOOST () 20 15 PHASE 10 5 0 1M
Gain will also increase with input voltage. The following graph shows the increase in GBW for an increase in supply voltage.
Bode Plot V =5V, V =1.8V, I
OUT
60 IN 50 40 GAIN (dB) 30 20
OUT
=3A
210 175 140 70 PHASE () 105 35 0 -35 -70 -105 1M
PHASE
GAIN (dB)
10 L=1H 0 COUT = 4.7F
GAIN
1k 10k 100k FREQUENCY (Hz)
-10 R1 = 10k R2 = 12.4k -20 C = 82pF FF -30 100 1k 10k 100k FREQUENCY (Hz)
The graph above shows the effects on the gain and phase of the system caused by feedback resistors and a feedforward capacitor. The maximum amount of phase boost achievable with a feedforward capacitor is graphed below.
5Vin, 1.8Vout at 3A load; * *
Phase Margin=43.1 Degrees GBW= 218KHz
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PHASE ()
105
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Max. Amount of Phase Boost Obtainable using CFF vs. Output 50 45 40 30 25 20 15 10 5 0 1 V
REF
MIC4723
Bode Plot V =3.3V, V =1.8V, I
60 50 40 GAIN (dB) 30 20 10 L=1H 0 COUT = 4.7F
IN OUT
Voltage
OUT
=3A
210 175 140 70 35 0 -35 -70 -105 1M PHASE () 105
PHASE
PAHSE BOOST ()
35
= 1V 5
2 3 4 OUTPUT VOLTAGE (V)
GAIN -10 R1 = 10k R2 = 12.4k -20 C = 0pF FF -30 100 1k 10k 100k FREQUENCY (Hz)
By looking at the graph, phase margin can be affected to a greater degree with higher output voltages. The next bode plot shows the phase margin of a 1.8V output at 3A without a feedforward capacitor.
As one can see, the typical phase margin, using the same resistor values as before without a feedforward capacitor results in 33.6 degrees of phase margin. Our prior measurement with a feedforward capacitor yielded a phase margin of 47 degrees. The feedforward capacitor has given us a phase boost of 13.4 degrees (47 degrees- 33.6 Degrees = 13.4 Degrees).
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MIC4723
dBm
Output Impedance and Transient Response
Output impedance, simply stated, is the amount of output voltage deviation vs. the load current deviation. The lower the output impedance, the better. Z OUT = VOUT IOUT
I =
10 10 x 1mW x 50 x 2 0.707 x R LOAD
The following graph shows output impedance vs frequency at 3A load current sweeping the AC current from 10Hz to 10MHz, at 1A peak to peak amplitude.
Output Impedance vs. Frequency
1 OUTPUT IMPEDANCE (Ohms) VOUT=1.8V L=1H =4.7F + 0.1 C
OUT
Output impedance for a buck regulator is the parallel impedance of the output capacitor and the MOSFET and inductor divided by the gain; R + DCR + X L Z TOTAL = DSON X COUT GAIN To measure output impedance vs. frequency, the load current must be load current must be swept across the frequencies measured, while the output voltage is monitored. Figure 9 shows a test set-up to measure output impedance from 10Hz to 1MHz using the MIC5190 high speed controller.
0.1 3.3VIN 0.01 5VIN
0.001 10
100 1k 10k 100k 1M FREQUENCY (Hz)
From this graph, one can see the effects of bandwidth and output capacitance. For frequencies <200KHz, the output impedance is dominated by the gain and inductance. For frequencies >200KHz, the output impedance is dominated by the capacitance. A good approximation for transient response can be calculated from determining the frequency of the load step in amps per second;
f=
A/sec 2
Figure 9. Output Impedance Measurement
Then, determine the output impedance by looking at the output impedance vs frequency graph. Then calculating the voltage deviation times the load step; VOUT = IOUT x Z OUT The output impedance graph shows the relationship between supply voltage and output impedance. This is caused by the lower Rdson of the high side MOSFET and the increase in gain with increased supply voltages. This explains why higher supply voltages have better transient response.
Z TOTAL = R DSON + DCR + X L GAIN X COUT
By setting up a network analyzer to sweep the feedback current, while monitoring the output of the voltage regulator and the voltage across the load resistance, output impedance is easily obtainable. To keep the current from being too high, a DC offset needs to be applied to the network analyzer's source signal. This can be done with an external supply and 50 resistor. Make sure that the currents are verified with an oscilloscope first, to ensure the integrity of the signal measurement. It is always a good idea to monitor the A and R measurements with a scope while you are sweeping it. To convert the network analyzer data from dBm to something more useful (such as peak to peak voltage and current in our case);
10
dBm 10
V =
x 1mW x 50 x 2 0.707
and peak to peak current;
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MIC4723
Ripple measurements
To properly measure ripple on either input or output of a switching regulator, a proper ring in tip measurement is required. Standard oscilloscope probes come with a grounding clip, or a long wire with an alligator clip. Unfortunately, for high frequency measurements, this ground clip can pick-up high frequency noise and erroneously inject it into the measured output ripple. The standard evaluation board accommodates a home made version by providing probe points for both the input and output supplies and their respective grounds. This requires the removing of the oscilloscope probe sheath and ground clip from a standard oscilloscope probe and wrapping a non-shielded bus wire around the oscilloscope probe. If there does not happen to be any non-shielded bus wire immediately available, the leads from axial resistors will work. By maintaining the shortest possible ground lengths on the oscilloscope probe, true ripple measurements can be obtained.
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MIC4723 Schematic and BOM for 3A Output
Item
C1a,C1b
Part Number
C2012JB0J106K GRM219R60J106KE19 08056D106MAT
Manufacturer
TDK Murata AVX AVX TDK Murata AVX Vishay VT Vishay Semi TDK Wurth Elektronik Vishay Dale Vishay Dale
Description
Qty
10F Ceramic Capacitor X5R 0805 6.3V
2
C2 C3
0402ZD104MAT C2012JB0J475K GRM188R60J475KE19 06036D475MAT
0.1F Ceramic Capacitor X5R 0402 10V
1
4.7F Ceramic Capacitor X5R 0603 6.3V
1
C4 D1 L1
VJ0403A820KXAA SSA33L RLF7030-1R0N6R4 744 778 9001 IHLP2525AH-01 1
82pF Ceramic Capacitor 0402 3A Schottky 30V SMA 1H Inductor 8.8m 7.1mm(L) x 6.8mm (W)x 3.2mm(H) 1H Inductor 12m 7.3mm(L)x7.3mm(W)x3.2mm(H) 1H Inductor 17.5m 6.47mm(L)x6.86mm(W)x1.8mm(H) 10K1% 0402 resistor 6.65k 1% 0402 For 2.5VOUT
1 1 1 1 1 1
R1,R4 R2
CRCW04021002F CRCW04026651F CRCW04021242F CRCW04022002F CRCW04024022F
Vishay Dale
12.4k 1% 0402 For 1.8 VOUT 20k 1% 0402 For 1.5 VOUT 40.2k 1% 0402 For 1.2 VOUT Open For 1.0 VOUT 1 1
R3
CRCW040210R0F
Vishay Dale
101% 0402 resistor
U1
Notes:
1. 2. 3. 4. 5.
MIC4723BML
Micrel, Inc.
3A 2MHz Integrated Switch Buck Regulator
1
TDK: www.tdk.com Murata: www.murata.com AVX: www.avx.com Vishay: www.vishay.com Wurth Elektronik: www.we-online.com
6.
Micrel, Inc: www.micrel.com
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MIC4723
Package Information
12-Pin 3mm x 3mm MLF(R) (ML)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2007 Micrel, Incorporated.
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